A synchronous buck-boost dc-dc controller, the Linear LT3791 and external MOSFETs can deliver 100W to power a string of LEDs (Fig. 1). The controller-MOSFET combination has an input range of 4.7V to 60V and an output range of 0V to 60V, along with a seamless transition between its buck and boost modes. These characteristics make it ideal for applications such as automotive, where the input voltage can vary dramatically during stop/start, cold crank and load dump scenarios, or portable battery operated systems where the battery output decreases with time. Transitions between buck, buck-boost, and boost operating modes are seamless, offering a regulated LED light output regardless of variations of supply voltage. Fig. 2 shows a typical application of the LT3791 driving a string of LEDs.
An internal differential current sense amplifier and resistor R1 monitor the input current and supply the sensed voltage drop between the IVINP and IVINN pins. When the voltage drop between IVINP and IVINN reaches 50mV, the controller goes into the constant-current mode. If the current sense voltage exceeds 50mV, it reduces the output current and regulates the current sense voltage to remain at 50mV.
The output differential current sense amplifier and sense resistor (R3) work like the input current sense but with a 100mV voltage drop between ISP and ISN. The output current sense level is also adjustable by the CTRL pin. An output current differential amplifier provides rail-to-rail operation. Similarly, if the FB (output feedback) pin goes above 1.2V it reduces the current level and regulates the output (constant-voltage mode).
As shown in Fig. 2, the LT3791 uses four external MOSFETs to provide from 5W to over 100W of continuous LED power with efficiencies up to 98.5%. LED current accuracy of +6% ensures constant lighting while + 2% output voltage accuracy enables the controller to operate as a constant voltage source. These N-channel power MOSFETs include two for the top switches (M1 and M4) and two for the bottom switches (M2 and M3).
The top MOSFET drivers are biased from floating bootstrap capacitors C1 and C2, which are normally recharged through an external diode when the top MOSFET is turned off. Schottky diodes across the synchronous switch M4 and synchronous switch M2 reduce the MOSFET’s voltage drop during the dead time. Inclusion of the Schottky diodes improve peak efficiency by 1% to 2% at a 500kHz switching frequency.
The LT3791 uses a proprietary technique to determine whether VIN>VOUT, VIN
When VIN>VOUT, the IC enters the buck mode where switch M4 is always on and switch M3 is always off. At the start of every cycle, synchronous switch M2 turns on first at which time the IC senses the inductor current. After the sensed inductor current falls below the reference voltage, synchronous switch M2 turns off and switch M1 turns on for the remainder of the cycle. Switches M1 and M2 alternate, behaving like a typical synchronous buck regulator. The duty cycle of switch M1 increases until it reaches the controller’s maximum duty cycle.
When VIN is close to VOUT, the controller is in buck-boost operation. Every cycle of controller turns on switches M2 and M4, then M1 and M4 until 180° later when switches M1 and M3 turn on, and then switches M1 and M4 turn on for the remainder of the cycle.
For low current operation, the LT3791 runs in forced continuous mode where the controller behaves as a continuous, PWM current mode synchronous switching regulator. In boost operation, switch M1 is always on, switch M3 and synchronous switch M4 are alternately turned on to maintain the output voltage independent of the direction of inductor current. In buck operation, synchronous switch M4 is always on, switch M1 and synchronous switch M2 are alternately turned on to maintain the output voltage independent of the direction of inductor current. In this mode, the output can source or sink current.
RT (Fig. 1) is the frequency adjust pin that allows programming of the switching frequency from 200kHz to 700kHz, which can optimize efficiency/performance or external component size. Higher frequency operation yields smaller component size but increases switching losses and gate driving current, and may not allow sufficiently high or low duty cycle operation. Lower frequency operation gives better performance at the cost of larger external component size. This requires an external resistor from the RT pin to GND.
You can synchronize the LT3791 to an external clock using the SYNC pin. Driving SYNC with a 50% duty cycle waveform is a good choice, otherwise maintain the duty cycle between 10% and 90%.
Power MOSFET Selection
The INTVCC pin is the output of an internal 5V regulator that sets the MOSFT drive voltage. Consequently, logic-level threshold MOSFETs must be used in LT3791 applications. If the input voltage is expected to drop below the 5V, then consider sub-logic threshold MOSFETs.
Power dissipated by the LT3791 must be known to select the appropriate power MOSFETs. For switch M1, the maximum power dissipation happens in boost operation when it remains on all the time. Its maximum power dissipation at maximum output current is:
RDS(ON) = On-resistance
ILED = LED current
VIN = Input voltage
VOUT = Output voltage
ρt= Normalization factor (unity at 25°C) accounting for the significant variation in on-resistance with temperature, typically 0.4%/°C. For a maximum junction temperature of 125°C, choose a value of ρt= 1.5.
Switch M2 operates in buck operation as the synchronous rectifier. Its power dissipation at maximum output current is:
Switch M3 operates in boost operation as the control switch. Its power dissipation at maximum current is:
Here, CRSS (reverse transfer capacitance) is usually specified by the MOSFET manufacturer. Constant k, which accounts for the loss caused by reverse-recovery current, is inversely proportional to the gate drive current and has an empirical value of 1.7.
For switch M4, the maximum power dissipation happens in boost operation, when its duty cycle is higher than 50%. Its maximum power dissipation at maximum output current is:
For the same output voltage and current, switch M1 has the highest power dissipation and switch M2 has the lowest power dissipation unless a short occurs at the output.
Important parameters for the four power MOSFETs are the breakdown voltage, VBR(DSS); threshold voltage, VGS(TH); on-resistance, RDS(ON); CRSS; and maximum drain current, IDS(MAX).
The operating frequency and inductor selection are interrelated because higher operating frequencies allow use of smaller value inductors and capacitors. Inductor value has a direct effect on ripple current. The maximum inductor current ripple is the maximum ripple that prevents sub-harmonic oscillation and also regulates the output with zero load. The ripple should be less than this to allow proper operation over all load currents. For a given ripple the inductance terms in continuous modes are:
f = Operating frequency
%Ripple = Allowable inductor current ripple
VIN(MIN) = Minimum input voltage
VIN(MAX) = Maximum input voltage
For high efficiency, choose an inductor with low core loss. To minimize its I2R losses, the inductor should also have low DC resistance. Plus, it must be able to handle the peak inductor current without saturating. To minimize radiated noise, use a shielded inductor.
Chose RSENSE based on the required output current. An internal current comparator threshold sets the peak of the inductor current in boost operation and the maximum inductor valley current in buck operation. The maximum current sensing RSENSE for boost operation is:
The minimum current sensing RSENSE for the buck mode is:
The final RSENSE value should be lower than the calculated RSENSE(MAX) in both the boost and buck operation. A 20% to 30% margin is usually recommended.
CIN and COUT
In boost mode, input current is continuous, whereas in buck mode, input current is discontinuous. Therefore, for the buck mode the selection of input capacitor, CIN, is based on the need to filter the input square wave current. Use a low ESR capacitor sized to handle the maximum RMS current. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life, so derate the capacitor.
In boost operation, the discontinuous current shifts from the input to the output, so COUT must be capable of reducing the output voltage ripple. The effects of ESR and the bulk capacitance must be considered when choosing the right capacitor for a given output ripple voltage.
To meet the ESR and RMS current handling requirements, it may be necessary to use multiple parallel capacitors. Ceramic output capacitors have good low ESR characteristics but can have a high voltage coefficient, so they are recommended for applications less than 100W. Capacitors available with low ESR and high ripple current ratings, such as OS-CON and POSCAP types may be needed for applications greater than 100W.
Programming LED Current
You program LED current by placing an appropriate value current sense resistor, R3, in series with the LED string. The voltage drop across R3 is (Kelvin) sensed by the ISP and ISN pins. Tie the CTRL pin to a voltage higher than 1.2V to get the full-scale 100mV (typical) threshold across the sense resistor.
The CTRL pin should not be left open (tie to VREF if not used). This pin can also be used with a thermistor for overtemperature protection of the LED load, or with a resistor divider to VIN to reduce output power and switching current when VIN is low. The presence of a time varying differential voltage signal (ripple) across ISP and ISN at the switching frequency is expected. The amplitude of this signal is increased by high LED load current, low switching frequency and/or a smaller value output filter capacitor. Some level of ripple signal is acceptable. Ripple voltage amplitude (peak-to-peak) in excess of 20mV should not cause operation problems, but may lead to noticeable offset between the average value and the user-programmed value.
There are two methods to control the current source for dimming. One uses the CTRL pin to adjust the current regulated in the LEDs. A second method uses the PWM pin to modulate the current source between zero and full current to achieve a precisely programmed average current. The minimum PWM on- or off-time is affected by choice of operating frequency and external component selection. The best overall combination of PWM and analog dimming capabilities occur if the minimum PWM pulse is at least six switching cycles and the PWM pulse is synchronized to the SYNC signal.
The LT3791 provides an open-drain status pin, SHORTLED*, which goes low when the FB pin is below 400mV. The only time the FB pin will be below 400mV is during start-up or if the LEDs are shorted. During start-up, the LT3791 ignores the voltage on the FB pin until the soft-start capacitor reaches 1.75V. To prevent false tripping after startup, a large enough soft-start capacitor (CSS) must be used to allow the output to get up to approximately 40% to 50% of its final value.
The LT3791 provides an open-drain status pin, OPENLED*, which goes low when the FB pin is above 1.15V and the voltage across V(ISP-ISN) is less than 10mV. If the open LED clamp voltage is programmed correctly using the FB pin, then the FB pin should never exceed 1.1V when the LEDs are connected. Therefore, the only way for the FB pin to exceed 1.15V is for an open LED event to occur.
Soft-start reduces the input power sources’ surge currents by gradually increasing the controller’s current limit. The soft-start interval is set by the soft-start capacitor (CSS). Besides soft-start, the SS pin is also used as a fault timer. If the controller detects an open LED or a shorted LED, it activates a 1.4µA pull-down current source. With a 100kΩ pull-up resistor to VREF on the SS pin, the LT3791 will continue to switch normally. With a 500kΩ pull-up resistor to VREF on the SS pin, the LT3791 will latch off until the EN/UVLO pin is toggled. Without any resistor to VREF the SS pin enters a hiccup mode operation. The 1.4µA pulls SS down until 0.2V is reached, at which point the 14µA pull-up current source turns on.
The LT3791 uses an internal transconductance error amplifier whose VC pin output compensates the control loop. The external inductor, output capacitor and the compensation resistor and capacitor determine loop stability.