An interesting magnetic design technique used by today's power supply engineers is the blending of transformer and inductive functions of dynamic power conversion circuits on a single magnetic core structure. When implemented correctly, this technique, known as integrated magnetics (IM) design, produces significant reduction of the magnetic core content of related power processing systems, resulting in cost-effective designs of lesser weight and smaller volume. Recently, IM packaging alternatives have been extended to include implementations in planar or flat forms, using modern printed-circuit approaches for windings together with low-profile ferrite core constructions.
It was demonstrated in 1984 ,  for circuit arrangements and in 1987  on a system level that all power conversion circuit designs of the switchmode variety have one or more IM forms. Prior to this, many believed only those power-processing networks wherein transformer and inductor winding potentials are always dynamically proportional could have IM versions . However, by using core structures that have more than one major material flux path , IM techniques can be applied to all switchmode power processing circuits and systems and can be extended to include other magnetic elements often excluded.
Let's look at two new approaches , ,  for using IM techniques in conjunction with planar core and winding construction methods. These approaches require the use of core structures with more than one winding window area and offer minimal interactions between the various magnetic elements included in the IM assemblies.
These approaches also reduce core system volume and weight by the IM process. The resulting core structures can enclose all windings (except for access slots), providing excellent magnetic shielding capability. In one of these approaches, off-the-shelf planar core sections can be used to construct the IM system.
Today, IM designs typically use soft-ferrite E-I or E-E core structures. Fig. 1 is a collection of traditional IM designs , ,  for a single-ended converter circuit, which differ only in winding locations on the three legs of the core structure. For the core leg where inductor windings are situated, an air gap is added to obtain the desired inductance values. The converter topology shown in Fig. 1(a), on page 22, is a buck-derived forward configuration, found in many dc power conversion systems where output power needs typically range from 50W to 250W.
Effective core leg areas must be chosen in accord with the maximum flux levels that occur as a result of converter operation so as to prevent saturation of any leg under maximum loading conditions of the system. As seen in the alternative designs in Fig. 1, each leg sees different flux levels, depending on winding locations and phase relationships, along with core leg reluctance values. For example, in the split-winding version shown in Fig. 1(c), the outer leg to the left of the center leg will see the sum of the transformer ac flux and one-half of the dc and ac flux produced by the inductor winding. In the right outer leg, the flux will be the difference between the transformer ac flux and the remaining half of the ac and dc components of the inductive flux generated by the winding(s) mounted on the center leg of the core structure. In this particular IM variation, the window area needs on each side of the center leg are equal. However, in the variation shown in Fig. 1(d), an optimum core design would require unequal window areas on each side of the center leg. This disparity is also a possibility for the IM designs in Figs. 1(b) and (e). In the variation of Fig. 1(e), the inductive leg is one of the outer legs of the core structure, and its area would need to be larger than the others, which is not a standard E-E or E-I core configuration. Finally, all of the alternative designs shown in Fig. 1 require winding bobbins that must be fitted on one or both of the outside core legs. Presently, such bobbins are nonstandard items, requiring custom manufacturing and added assembly cost.
For push-pull versions of converter circuits where two-quadrant B-H loop operations occur from transformer actions, the center-most leg is used for the inductive part of the converter, with dual primary and secondary windings placed on the outer core legs. In these situations, the window area requirement is balanced, such as the core-and-winding arrangement shown in Fig. 1(c).
A variety of methods for core reset  due to the single-ended transformer action of the converter in Fig. 1(a) are available. Some of these methods use resonant reset techniques involving the parasitic capacitances of T1, Q1, D1, and D2 and self-inductances of the transformer windings.
Planar IM Concepts
Using printed wiring methods for the windings of an IM can lower the height profile of the overall magnetic component package. Windings are mounted on a laminate base, eliminating the cost of special bobbins. Fig. 2, on page 26, shows a Planar Integrated Magnetic (PIM) design for a split-winding version of the forward converter topology.
In Fig. 2, the primary and secondary windings are interleaved, using selected layers of the PCB on the outside portions, while the inductor winding is split into parallel sections of the inside layers. Feed-through vias in the PCB are used to interconnect the various parts of all three windings. Terminations of the winding ends can be accomplished in several ways, including the use of solid pins running vertically from the PCB end patterns down to the PCB of the converter assembly. The use of gull-wing leads is another alternative, which is useful when placing the PIM on a motherboard assembly with surface-mounted components.
Conventional IM constructions, whether using PCB-style or wire windings, have limitations. Because windings are required on the outer legs of the core system, it isn't possible to completely surround all windings with core material to restrict magnetic leakage levels. This situation is seen from the PIM construction example shown in Fig. 2. Also, because conventional constructions using E-I or E-E cores restrict window locations for windings to two locales and material flux paths to a maximum of three, other power magnetic components in a conversion system (such as input and added second-stage output filter inductances) can't be easily accommodated in an IM arrangement without significant topology changes. This, in turn, often leads to compromises in power processing performance.
Rather than placing the various windings of a PIM in a conventional open, side-by-side manner, you can arrange them in a closed, top-to-bottom core configuration , , as shown in the cross-sectional diagram in Fig. 3(b). Here, the upper chamber of the core serves as the location for the output inductive windings of the converter, while the lower chamber is for the windings of the transformer. The core piece separating the two chambers provides a material path for inductive and transformer flux elements. With the windings phased as shown in Fig. 3(b), the effective flux level in this common path is the difference between these flux values. Therefore, the core material needed for this common path is reduced accordingly. Although the center post areas shown in Fig. 3(b) are identical, this condition isn't an absolute design requirement. Optimum size and unit height studies for the PIM form shown in Fig. 3(b) for a particular converter application may indicate otherwise.
Fig. 4. shows a practical construction of the PIM concept of Fig. 3(b). Here, a pot format is used, consisting of three separate core pieces stacked to form the two window areas, or winding chambers, needed for the transformer and inductor parts of the IM. Within the lower chamber is the PCB assembly for the primary and secondary windings, with the upper chamber reserved for the PCB assembly for the inductor winding. It's possible to use double-sided PCBs in place of a single multilayer PCB in each locale, and interconnect them by external wiring methods using termination tabs on the PCBs. These tab areas are accessed by means of slots cut into the core sections, as seen in Fig. 4.
In this construction example, a small hole through the center posts of the cores permits a single nonconductive screw-and-nut combination to hold the assembly together.
Simply adding one or more inductive windings to a stacked PIM construction to accommodate additional filter inductances of a converter system is a matter of adding a third chamber area ,  to the core arrangement. This chamber can be located below the chamber in Fig. 3(b), on page 27, where the transformer windings are housed. Fig. 5(b) illustrates this construction, wherein the lower chamber contains the winding(s) for the inductance of an input filter network for the forward converter topology shown in Fig. 1(a), on page 22. Now, the core piece below the transformer chamber serves as a differential flux path for the transformer ac flux and the dc + ac flux generated by the lower winding associated with the input filter inductance.
Because the reluctance of the upper and lower air gaps is larger than those posed by the inner and outer material paths of the core structure, flux produced by the MMF of the transformer section of the PIM is restricted to inside sections of the core system. Thus, PIM inductive and transformer operations are independent, with insignificant magnetic interaction. This situation implies that there are no significant restrictions on turn ratios between transformer and inductor windings, whereas in some designs , such restrictions are required to ensure proportionality of winding driving potentials to permit their magnetic components to use a common core structure.
In the cross-sectional view of the PIM construction in Fig. 5(b), on page 28, the unshaded areas on the outside walls of the core structure represent open parts of the core for accessing the ends of the PCB windings located in each chamber. Fig. 6 shows construction approaches for viable PIM assemblies of the stacked variety , , with typical access locations for PCB winding terminations illustrated.
Another PIM construction alternative is a side-by-side arrangement of transformer and inductive windings , as depicted in Fig. 7, on page 31. In contrast to Fig. 3, on page 27, the inductive portion of the PIM lies in the center of the structure, with the transformer windings in a second chamber that surrounds the center portion. The core wall that separates the two chambers of the core system then serves as a common flux path for inductive and transformer operations. Fig. 8, on page 31, is a sketch of a practical implementation of the PIM concept  of Fig. 7. A cross-sectional view of the design is shown in Fig. 9, on page 32.
As indicated in Fig. 9, interactions between the magnetic operations of the system can be minimized further by the presence of a small air chamber placed in this common flux path. It's also feasible to place a shield band of conductive material in this chamber. This shield technique is similar to a bellyband copper screen often added around the outside of a conventional inductor or transformer to reduce radiated magnetic emissions.
The PIM method shown in Fig. 7 keeps the core structure height low. However, the surface area of the PIM increases over the stacked arrangement of Fig. 3, requiring more area for mounting. To access the inductive winding, holes must be placed in the bottom part of the inner core chamber. It's possible to add a third outer chamber and associated core walls to the outside part of this core system for another set of inductive windings. However, because the winding lengths are longer than those of the innermost chambers, copper losses will be higher than those in the 3-chamber stacked design approach illustrated in Fig. 5.
Many power converter systems require multiple inputs or outputs that, in turn, require additional inductances for filtering purposes. These inductances can be placed in a PIM in coupled-inductor arrangements ,  in the appropriate chambers. For example, in the PIM design shown in Fig. 5, leakage inductance values between windings mounted in the upper or lower chambers can be controlled by the addition of thin disks of core material . Such disks provide magnetic control and reduction of ac current levels in selected inductor windings , , . Fig. 10, on page 32, shows examples of reluctance disk forms. Suitable disk materials include inexpensive varieties of cold-rolled steel and low-permeability soft ferrite. Disks of nonmagnetic materials can also be used where desired leakage inductance values needed are small.
PIM Modeling Methods
To better understand the magnetic operations of the PIM designs shown in Fig. 5 and Fig. 7, equivalent circuit models can be developed using the reluctance-to-inductance modeling methods described in Chapter 12 of . These models can then be used to study the dynamics and magnetic interactions between the transformer and inductive sections of a PIM.
For example, using the flux paths and directions defined earlier in Fig. 6 for the 3-chamber PIM structure of Fig. 5, a first-order reluctance model of the magnetic system is formed, along with MMF sources. This model is illustrated in Fig. 11, on page 32. Note that the symmetry of the basic model permits it to be simplified as indicated in this sketch.
With a base reluctance/MMF model established, it can be converted into one involving inductances and excitation sources. This new model is shown in simplified form in Fig. 12, on page 32. In this model, all inductance values are referred to winding NP of the PIM system in Fig. 5. Values for the inductances indicated in Fig. 12 can be estimated using the first-order reluctance and inductance relationships defined in the table.
Examination of the resultant circuit model in Fig. 12 shows that the two inductances (Lit and Lib) formed by the two core pieces separating the two inductive chambers from the transformer chamber will be much larger in value than those associated with the upper and lower core pieces where air gaps are present (Lct and Lcb). For this reason, very little of the flux developed by the transformer actions within the center chamber will appear in the core areas associated with the inductor chambers. This confirms that transformer and inductive operations in this PIM will be largely independent, with very little interaction between them.
Finally, as an example of the use of this circuit model in analyzing its use in conjunction with a converter network, Fig. 13 is a circuit diagram of the forward converter system of Fig. 5, redrawn to include the PIM model of Fig. 12.
PIM Prototype Tests
To verify the stacked PIM approach illustrated in Fig. 5, a 200 kHz, 20Vdc to 40Vdc in, 5V 10A out PIM forward converter system was designed, built, and tested for performance, as a part of a recent NASA SBIR Phase II proposal effort. In this case, a 3-chambered PIM was constructed by using four separate cylindrical pieces of soft ferrite of the MN8CX variety made by Ceramic Magnetics. Overall height of the PIM structure was 16.8 mm (0.661 in.), and its diameter was 35.2 mm (1.386 in.). Center post area in all chambers was set by design to 53.5 mm2 (0.076 in.2), with air gap lengths in the upper and lower chamber center posts cut to 254 µm (0.01 in.). The primary and input inductor windings used three paralleled 8-turn double-sided PCBs, while the secondary and output inductor windings used three paralleled double-sided 5-turn PCBs. Four-ounce copper patterns were used for all PIM PCB windings.
Measured inductance of the primary winding, the input filter inductor and the output filter inductor was 250 µH, 19.2 µH, and 7.5 µH, respectively — very close to design projections. The measured efficiency of the power stage under maximum loading conditions was 87%, with total PIM core and winding power losses measured at nominally 1.2W. The measured temperature rise above ambient of the PIM was 30°C (no heat sink or forced-air cooling). As predicted by design, no discernible magnetic interactions were observed with regard to the transformer and inductive functions within the PIM during the testing of the converter. Core volume and weight savings over a conventional converter design having two individual planar inductors and one planar transformer were 29.5%.
Similar verification testing of the side-by-side PIM system (shown in Figs. 8, page 31, and 9, page 32, have been reported by research engineers in Japan ,  with equal success. In one experiment, an off-line 100Vac to 24Vdc, 125W, 350 kHz forward converter system was built and tested. The PIM was constructed using a high-frequency low-loss ferrite of the 2500B2 variety made by TOKIN (Sendai, Japan). The outside diameter of the PIM was 53 mm (2.09 in.), and its height was 8 mm (0.315 in.). Total center post gap length was set at nominally 300 µm (0.012 in.) to yield an output filter inductance of 16 µH. A 5-turn primary winding and a 3-turn secondary winding were used in the outer chamber, with an inner 6-turn inductor winding. Two-ounce copper patterns, nominally 70 µm (0.0028 in) thick, were used for all multilayer windings to minimize high-frequency copper losses.
The efficiency of the PIM of this converter system was measured to be on the order of 98% at an output power level of 125W. Noise reduction tests showed 10 dB to 20 dB reductions in 100 MHz to 300 MHz radiated noise over that of a comparable converter design with separate open-frame inductor and transformer elements.
These top-to-bottom and side-by-side multichambered PIM constructions can include simple planar disks of magnetic material to control leakage inductances in selected areas of the structures. Form factors for the top-to-bottom construction variations can be either open or closed. In the former case, these PIMs can use standard circular core halves (e.g. pot, RM, PQ, DS shapes), or multiple E-I combinations of off-the-shelf low-profile rectangular cores.
The author acknowledges the funding support provided by the U. S. National Aeronautics & Space Administration (NASA) in the early phase of development  of the stacked PIM concepts described in this paper. A U. S. Patent  was issued to the author's company in 1998 relative to this PIM method and other related enhancements.
The author also acknowledges the work performed by Dr. Toru Fujiwara and his associates in the independent development of the complementary side-by-side PIM structure ,  shown in this paper. For more information, contact him at the National Matsushita Electric Works Ltd. in Kadoma, Osaka, Japan, or at [email protected].
R. P. Severns and G. E. Bloom, Modern DC-to-DC Switchmode Power Converter Circuits, Reprint Edition, e/j BLOOM associates Inc., 1990, pp. 262-324 (ISBN 0-442-21396-4, Van Nostrand Reinhold, 1985).
S. Cuk, “A New Zero-Ripple Switching DC-to-DC Converter and Integrated Magnetics,” IEEE Power Electronics Specialists Conference Record, 1980, pp. 12-32 (IEEE Pub. 80CH1529-7).
G. E. Bloom, “Multi-Chambered Planar Magnetics Design Techniques,” IEEE Power Electronics Specialists Conference Record, 2000, pp. 295-301 (IEEE Pub. 00CH3701-8).
G. E. Bloom and R. P. Severns, “The Generalized Use of Integrated Magnetics and Zero-Ripple Techniques in Switchmode Power Converters,” IEEE Power Electronics Specialists Conference Record, 1984, pp. 15-33 (IEEE Pub. 84CH2000-8).
G. E. Bloom, “New Integrated-Magnetic Power Converter Circuits for Telecommunications Systems,” IEEE INTELEC Conference Record, 1984, pp. 359-366 (IEEE Pub. 84CH2073-5).
G. E. Bloom, “New Integrated-Magnetic DC-DC Power Converter Circuits and Systems,” IEEE APEC Record, 1987, pp. 57-66 (IEEE Pub. 86CH2312-7).
G. E. Bloom, “Integrated Magnetic Apparatus,” U. S. Patent 5,726,615 (March 10, 1998).
G. E. Bloom, “New Multi-Chambered Power Magnetics Concepts,” IEEE Trans. Magnetics (1997 INTERMAG Conference Issue), July 1998, Vol. 34, No. 4, pp. 1342-1344 (IEEE Pub. ISSN 0018-9464).
T. Fujiwara, “Planar Integrated Magnetic Component with Transformer and Inductor using Multilayer Printed Wiring Board,” IEEE Trans. Magnetics (1997 INTERMAG Conference Issue), July 1998, Vol. 34, No. 4, pp. 2051-2053 (IEEE Pub. ISSN 0018-9464).
T. Fujiwara, “Analysis of Iron Loss of Planar Integrated Magnetic Component with a Transformer and Inductor,” Trans. Magn. Society, Japan, Vol. 23, No. 4-2, 1999, pp. 1629-1632 (ISSN 0285-0192).
J. West and L. Dickstein, “New Common-Mode Choke Structure for Switch Mode Power Supplies,” POWERTECHNICS Magazine, Nov. 1985, pp. 29-31.
G. E. Bloom, Planar Integrated-Magnetic Power Components (Phase One Studies), NASA SBIR Final Report NAS7-1225, Aug. 13, 1993.
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