Spurious electromagnetic energy pollutes the local environment near a noise-generating source, such as an ac-dc power supply or dc-dc converter. This electromagnetic energy generates both electric and magnetic fields, which can propagate into the environment in one of two ways: via the high frequency noise currents that flow through the interfacing lines (conducted emissions), or radiation through the atmosphere surrounding the noise generating source (radiated emissions). Excess amounts of either form of EMI can cause disturbances and/or malfunction of electronic equipment within the proximity of the noise-generating source.
No one can completely eliminate EMI, but it is possible to suitably attenuate “safe” levels to ensure reliable operation of a system and to help achieve that goal international regulatory agencies have recommended acceptable emission levels. These agencies include CISPR (International Special Committee on Radio Interference), FCC and VDE.
AC-DC power supplies and dc-dc converters generate noise because they switch current at a fundamental high frequency, relying upon fast turn-on and turn-off of the main switching element — invariably a power MOSFET device. This results in fast rise and fall time high frequency current pulses containing significant amounts of sub-harmonic energy. The fundamental switching frequency of the majority of today commercially available switching power supplies lie within the 200 kHz to 1 MHz range, whereas the sub-harmonic content of these fundamental frequencies can extend into the 10 MHz range.
Such switching power supplies generate conducted EMI consisting of differential-mode and common-mode elements. Differential-mode currents flow in and out of the switching power supply via the power leads and its source (or load), and is totally independent of any grounding arrangement. Consequently, no differential-mode current flows through the ground connection. On the other hand, common-mode currents flow in the same direction either in or out of the switching power supply via the power leads and return to their source through the lowest available impedance path, which is invariably the ground connection. Even if the ground connection is not deliberate, common-mode currents flow through the capacitance between the case of a system and its ground.
At frequencies below approximately 5 MHz, the noise currents tend to be predominantly differential-mode, whereas at frequencies above 5 MHz, the noise currents tend to be predominantly common-mode. Furthermore, conducted EMI emissions are generally only measured up to 30 MHz.
Differential-mode currents are effectively attenuated by connecting bypass capacitors directly between the power lines of the switching power supply. These power lines may either be at the input, and/or at the output, of the switching power supply. Ideally these bypass capacitors need to be physically located immediately adjacent to the terminals of the noise-generating source to be most efficient.
Conventional p. c. board traces exhibit an inductance of approximately 10nH per inch, which appears in series with any bypass capacitor connected in this fashion. Moreover, this inductance appears in both the positive power feed path, as well as its power feed return path, implying that any bypass capacitor physically located one inch away from the input terminals of the noise generating source exhibits approximately 20nH of series inductance. This series inductance together with the bypass capacitor form a resonant network, which appears as a capacitor below resonance, and an inductor above resonance. The effectiveness of the bypass capacitor is seriously impaired as the sub-harmonic frequencies of a switching power supply exceed this resonant network frequency.
Consider a typical example of a 1µF ceramic capacitor physically located just one inch away from a noise-generating source. Assuming the series inductance given previously, the resonant frequency of this bypass capacitor network is only 1.1 MHz, implying that any sub-harmonic frequencies in excess of this resonant frequency will not be as efficiently attenuated. As previously stated, differential-mode sub-harmonic frequencies are typically present up to approximately 2 MHz, at which point this fairly typical bypass capacitor network is already seriously degraded. Remember that a switching power supply with a 400 kHz oscillator frequency will have its third harmonic content apparent at 1.2 MHz. The arrangement described immediately above is a fairly typical differential-mode filtering arrangement applied to switching power supplies, both in terms of the values selected and the distances of p. c. board traces.
As we progressively increase the distance of this 1µF ceramic capacitor away from the noise generating source the series inductance proportionally increases, forcing the resonant frequency of the bypass capacitor network to further reduce, causing even more degradation in the filtering performance of this bypass capacitor. Consequently, the actual location of the bypass capacitor is critical for efficient attenuation of differential-mode currents at high frequencies.
Attenuation at lower frequencies of differential-mode currents, in or around the fundamental switching frequency of the noise generating source, may dictate that a much higher value of bypass capacitance is required that cannot normally be attained with a ceramic style capacitor. Current ceramic capacitors are only available up to approximately 22uF at low voltage ratings, which maybe suitable for effective differential-mode filtering across the lower voltage outputs of switching power supplies, but not suitable across the inputs of switching power supplies where 100V surges can often be experienced. For these applications electrolytic style capacitors are often employed because of their high capacitance and voltage ratings, however this style of capacitance also exhibits a high value of ESR. A typical range of ESR values for an electrolytic style capacitor is 200mW to 500mW, which appears as a series impedance to the filter capacitance in much the same way as the series inductance previously described does, thus degrading the filtering performance of the bypass capacitor particularly at higher frequencies. For comparison purposes, a typical ESR value for a ceramic capacitor may only be 10mW, making them far more suitable for higher frequency attenuation of switching frequency sub-harmonic components. Consequently, many differential-mode input filters consist of a combination of electrolytic and ceramic capacitors to suitably attenuate differential-mode noise currents both at the lower fundamental switching frequency as well as at the higher sub-harmonic frequencies.
Take care when placing filter capacitors, either at the input or at the output, of a switching power supply to ensure that the current flow actually passes across the terminals of the filter capacitor for most effective attenuation. Use of four-terminal style techniques for filter capacitors should be practiced, where the p. c. board trace enters the filter capacitor node upon one side of its surface-mount (or through-hole) pad, and departs from the opposite side of the same pad. Filter capacitors should never be “spurred” off from the main current carrying p. c. board trace.
Further attenuation of differential-mode currents can be achieved by adding an inductor in series with the main power feed to form a single stage LC differential-mode low-pass filter with the bypass capacitor(s), as previously described. The combination of inductor, L, and bypass capacitor, C, attenuate higher frequency differential-mode currents more effectively while at the same time allow low frequency or dc operating currents to pass through unaffected.
Some knowledge of the operating or switching frequency of the noise-generating source is required to calculate the cut-off characteristics for this low-pass filter. Furthermore, as switching power supplies exhibit a negative input impedance, exercise caution in selecting the values of inductor and bypass capacitor for this low-pass filter. Mismatches between the output impedance of the low-pass filter and the input impedance of the switching power supply can cause serious operational instabilities. Normally, the location of a large enough bypass capacitor physically located close to the input terminals of the switching power supply ensures this instability condition is not attained because it lowers the bus impedance as seen from the input terminals of the switching power supply.
Fig. 1 is a typical example of an effective single stage LC differential-mode low-pass filter as detailed above. Note also that for particularly harsh EMI environments additional LC stages could precede the single stage LC differential-mode low-pass filter illustrated in Fig. 1, providing a multi-stage low-pass filter that offers significantly more attenuation to differential-mode currents.
Common-mode currents are effectively attenuated by connecting bypass capacitors between each power line of the switching power supply and system ground. These power lines may either be at the input and/or at the output of the switching power supply. To be most efficient, these bypass capacitors must be physically located adjacent to the terminals of the noise-generating source.
The task facing the system designer is to provide the shortest possible path for the return of common-mode noise currents, remembering that these currents always flow through the lowest available impedance path. As previously noted, common-mode currents are invariably apparent above 2 MHz, and still prevalent well beyond 30 MHz. Unfortunately, for common-mode currents in this frequency range it is often difficult to predict which path that actually will be. However, in general reducing the inductance and increasing the capacitance of the loop through which the common-mode currents flow will reduce the overall impedance of that loop. Because the inductance of a wire loop is proportional to the area of that same loop, the smaller the loop area the lower its inductance and, as a consequence, the higher the capacitance and the lower the overall loop impedance. Because of the frequency range of the common-mode currents previously described, very low ESR ceramic bypass capacitors are the most suitable option for effective common-mode filtering. However, exercise care when applying a common-mode filter capacitor arrangement to an isolated power supply.
Invariably the input-to-output isolation breakdown voltage exceeds 1.5kV. This implies that if the system ground point were referenced to the input side of the power supply being filtered then any common-mode bypass capacitor fitted to the output side of same power supply would have to maintain this breakdown voltage. Criteria for system safety agency approval must also be maintained to adhere to safety requirements such as Basic Isolation, Functional Isolation, etc. Y-Class capacitors are commercially available that exhibit self-healing properties that support this application, but capacitance values are relatively low, often failing to suitably attenuate the common-mode EMI by themselves.
However, further attenuation of common-mode currents can be achieved by adding a pair of coupled choke inductors in series with each main power feed. The coupled choke inductors must be wound on a single magnetic structure, such as a toroidal core, with winding polarities that ensure flux cancellation of low frequency dc currents to prevent core saturation. In this arrangement, the high impedance of the coupled choke inductors at high frequencies block the exit of common-mode currents to the outside world, forcing the same common-mode currents through bypass capacitors described previously. Remember that the impedance of an inductor increases with frequency, whereas the impedance of a capacitor decreases with frequency. Fig. 2 is an example of an effective common-mode filter applied to a typical system installation of equipment incorporating a noise-generating source as detailed above.
Switching power supplies also generate radiated EMI emissions. Radiated EMI appears in the form of electromagnetic waves that “radiate” into the immediate atmosphere directly from the circuitry and its interface leads. The circuitry and its interface leads can liken themselves to a transmitting antenna for this radiated EMI. Radiated EMI emissions are generally measured at much higher frequencies than their conducted counterparts, namely beyond 30 MHz up to several GHz.
Radiated EMI can contain electric and magnetic fields. The strength of the electric field is proportional to the circuit voltage, operational frequency and “the effective length of the antenna.” The strength of the magnetic field is proportional to the circuit current, operational frequency and “the effective area of the antenna loop.” Because the circuit parameters and operational frequency are fixed for a systems' operating characteristics, the only variable is the length of, or the enclosed loop area formed by, the power line its return path. Therefore, it can be seen that radiated EMI can be minimized by physically locating the noise-generating source, or switching power supply, as close to its source and load as is physically possible. However, system mechanics rarely permit such a compact assembly to be achieved, and in most cases the switching power supply is located some distance from one, or even both, of the above.
Even within such systems though, the reduction in the length of, or the enclosed loop area formed by, the power line and its return path can still be designed to efficiently minimize radiated EMI. The inductance of a p. c. board trace can be minimized by making it as wide as possible and routing it parallel to its return path. Similarly, because the impedance of a wire loop is proportional to its area, reducing the area enclosed between the power line and its return path will further reduce its impedance. Within p. c. boards this area can be best reduced by tracking the power line and its return path one above the other upon adjacent printed circuit board layers. Remember that by reducing the loop area between a power line and its return path not only reduces the RF impedance, but also reduces the efficiency of the antenna because its smaller loop area generates a lower magnetic field. Refer to Figs. 3(a) and 3(b), where the arrangement illustrated in Fig. 3(b) is far superior to that presented in Fig. 3(a) in terms of radiating EMI.
Ground planes located upon the outer surfaces of the printed circuit board, particularly directly below the noise-generating source is strongly recommended and significantly reduces radiated EMI. Attempts should be made to extend this ground plane beyond the edges of the noise-generating source. Where a noise generating source has an outer case or baseplate surface similar to those apparent upon many “brick” style switching power supplies, attempts should be made to form a connection between this surface and the p. c. board ground plane. This would provide a low impedance path for the return of common-mode currents. You should also avoid routing noise sensitive p. c. board signal traces underneath noise generating sources.
Metal shielding is another recognized practice for suppressing radiated EMI, where the noise generating source is physically located inside a grounded conductive housing. Interface to the “clean” outside environment maybe achieved via in-line filters or filter connectors. Common-mode bypass capacitors would need to be returned to a ground bond star-point located somewhere upon this conductive housing structure.
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