A 48kW resonant converter includes four power modules, each containing two paralleled IGBTs and antiparallel diodes. They are arranged in a half-bridge or push-pull configuration depending on the input, at 400Vac or 200Vac. At maximum power, peak load current is 550A at 50 kHz, or 275A per module for 48kW out. The generator is zero-voltage switched to create a continuous series resonant output current that's transformer-isolated — stepped up and rectified to the desired output level. The output voltage is regulated by a DSP-based frequency-modulated controller, with dual loop feedback on resonant current and load kV.
For a range of output power, the system operates from 48 kHz up to 68 kHz with a resonant LC shunt across the load transformer. With fundamental series resonance at 48 kHz, the shunt resonates at 68 kHz. At low frequencies, the generator functions near resonance, with high power throughput. As frequency increases, the impedance rises — the load being shorted by the resonant shunt. At 68 kHz, power is zero. Fig. 1 shows the primary-side power circuit.
Minimum size is crucial for state-of-the-art X-ray equipment. In this version of the converter, the four ZVS modules with their tightly packed IGBT and FRED chips require only ¼ of the surface area formerly used. Integration of drivers, isolation, and ZVS logic circuitry further shrinks the footprint. Control signals have less distance to travel, which improves noise immunity and mechanical integrity.
However, there's a price to pay for compactness. With power semiconductors now more concentrated, heat density is higher and thermal management exacerbated. Maintaining the heatsink at 50°C requires a substantial 50 cm × 40 cm × 10 cm aluminium blown-air cooled extrusion. The use of direct-bonded-copper (DBC) aluminium nitride substrates, soldered onto state-of-the-art metal-matrix (AlSiC) baseplates, optimizes thermal management inside the modules. There's also a liquid-cooled version of the module in which the water chamber doubles as a module baseplate, resulting in a very low chip-to-coolant thermal resistance. Its total size is about that of the air-cooled module minus the heatsink.
The advent of IGBTs and power MOSFETs has created new opportunities for advanced converter topologies. In particular, detailed analysis of commutation mechanisms has led to greater recognition of the merits of soft switching, with its improved performance at high frequencies. One intriguing topology is the “thyristor-dual,” an artificially-synthesized switch (Fig. 2, on page 26) combining the loss-free turn-on of a diode with the gate-controlled turn-off of a real switch. Thyristor-duals are generally configured as phase legs for use in voltage-fed inverters. They're created by connecting a semiconductor switch that can be turned off (BJT, IGBT, MOSFET, etc.) in antiparallel with a diode and snubber capacitor. An AND gate allows turn-on when zero-voltage and turn-on validation concur. In Fig. 2 and Fig. 3, on page 27, subscripts 1 and 2 define the switches and their variables.
In Fig. 3, VK1 is the voltage across K1, ICH the alternating current to the load. K1 and K2 are switch gate signals. For simplicity, ICH is shown as a sine wave; however, this isn't restrictive since only the position of its zero points with respect to the gate signals is important.
Starting from t° in Fig. 3, the following sequence occurs:
- At t = t° +: VK1 = E and VK2 = 0, corresponding to K1 OFF and K2 ON.
- With ICH positive, D2 conducts.
- Because VK2 = 0, validating T2.
- At t = t1: ICH changes sign and T2 conducts.
- At t = t2: T2 turns off, ICH transfers to C1/C2, VK2 rises and VK1 falls.
- At t = t3: D1 conducts, validating VK1 = 0 and T1.
- At t = t4: ICH changes sign and T1 conducts.
- The sequence completes when T1 transfers to D2 in the same manner as T2 to D1.
This mode always prevents shoot-through in the branch. First, the circuit validates the turn-on signal to any one switch, but only when the voltage across its complement is E. This implies the latter must be solidly “off” before the former can turn on. Second, the provision of snubbers minimizes dv/dt. Excessive reapplied dv/dt following device turn off can provoke shoot-through, as the Miller-coupled charge tries to turn the device back on again. This logic makes the need for dead time unnecessary.
Load current charges the capacitors as the switch voltage rises. Energy exchange accompanies turn off between load and capacitors. In a dc chopper or PWM inverter, this energy is lost as heat in the RCD snubber during switch turn-on. Current flows in the capacitors as the switch voltage collapses in the thyristor-dual load, so the energy returns to the load. Because such a snubber is virtually loss-free, you can increase its capacitance value to minimize turnoff losses in the switch. In this configuration, turn-on losses are negligible, thanks to the spontaneous turn-on.
This system employs a 1200V/400A power module for use in this 48kW converter (Fig. 4). Configured as a single switch to operate in the ZVS dual-thyristor mode, each module contains 1200V rated IGBTs and matched antiparallel diodes, as well as logic circuitry necessary to provide full ZVS functionality. Two modules are paralleled for the high side and two for the low side of the half-bridge phase leg.
This module contains an IGBT driver and control logic circuitry, along with a high-frequency transformer isolated internal ±15V power supply, and undervoltage-lockout circuits. A four-layer p. c. board positioned close to the power switches holds the circuits. A layer is a ground plane for enhanced noise immunity. Circuit packaging emphasizes cooling. The high-peak current driver output stages — implemented with SMDs — are mounted on a small alumina substrate soldered to the metal-matrix AlSiC baseplate. An NTC thermistor provides over-temperature feedback.
Galvanically-isolated control signals employ the high-frequency power supply transformer, which enhances electrical performance and boosts reliability. CMOS control-logic compatibility requires a single 12V supply on the primary side. Special start-up circuitry ensures that the system will power-up properly with applied voltage.
While the efficiency of a ZVS converter like this is unmatched at the high frequencies employed, dissipation in the modules is still substantial, given their compact nature and the high power throughput. For maximum throughput at reasonable junction temperatures, all power semiconductors are mounted on aluminium nitride substrates for highest thermal conductivity.
Power connections are M5 screws, with logic level and auxiliary supply circuits interfaced through 0.6 mm × 0.6 mm pins on a 0.1 in. raster, allowing direct mounting of the control board without wire links. In this way, parasitics are minimized and reproducibility assured.
Switching losses are less than 1%, with output regulation less than 1% over the range 80kV to 140kV. The ripple is an incredibly low 180V peak-to-peak below 1 kHz. Because of the need for small size in an X-ray converter, the application-specific power modules themselves are very compact with the smallest possible footprint.
- H.Foch, P.Marty, J.Roux, “The Use of Duality Rules in the Conception of Transistorized Converters,” PCI Proceedings, Munich 1980, pp. 4B3 1-11.
- Y.Cheron, H.Foch, J.Roux, “Power Transfer Control Methods in High Frequency Resonant Converters,” PCI Proceedings, Munich, 1986, pp 92-103.
- Y.Cheron, Lavoisier, “La Commutation Douce,” TEC et DOC, France, 1989.
- M.Metz, J.P.Arches, H.Foch, S.Boyer, D.Ferrer and S.Ben Doua, “Switching limits in Static Converters Using Soft Commutation,” Power Electronics and Variable Speed Drives, IEE, London, July 1990.
- H.Foch, M.Metz, JP.Arches, Ch.Saubion, “Les Composants à Grille Isolée en Mode Thyristor-Dual,” Revue Générale d'Electricité, Février 1994, No. 2, p. 13-22.
- Serge Bontemps, Alain Calmels, MOS-gated power semiconductors configured in the ZVT thyristor-dual mode yield >95% converter efficiency at 1kW to 10kW, when resonantly switched at 20 kHz to 400 kHz.
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