In the conventional design of a forward converter, the maximum duty cycle can be extended above 50% by using a reset and clamp technique. This technique, known as RCD clamp circuitry, consists of a resistor, capacitor, and diode. These components are used to develop a varying clamp voltage into which the magnetizing energy is discharged and dissipated as heat.
Compared to the conventional forward converter design, the RCD clamp eliminates the primary reset winding, reducing the transformer production costs. The RCD clamp technique is commonly used in designs where a wide input voltage range requires management. In particular, designs with low input voltage ranges benefit significantly from this technique, since the component stresses are still manageable with low voltage semiconductors. Despite the benefits of the RCD clamp, the switching FET has to withstand a high voltage stress during the reset of the RCD. Additionally, switching losses in the main FET can significantly reduce the overall efficiency and limit the operating frequency of the converter. Fig. 1 shows the basic schematic of an RCD forward converter.
Forward Converter with Active Clamp
In an active clamp design, a power MOSFET replaces the resistor and diode of an RCD clamp. The same IC that controls the main power MOSFET controls the clamp MOSFET. You can see a simplified schematic of a forward converter with an active clamp and self-driven synchronous rectifiers in Fig. 2.
By using an active clamp technique instead of the conventional RCD clamp, the efficiency of converter is increased in several ways. First, the active clamp provides soft switching for both the main FET and clamp FET. In addition, implementing synchronous rectifiers on the secondary side provides a significant boost in efficiency. Finally, instead of dissipating the magnetizing energy in a clamp resistor, the magnetizing energy is recycled back to the input source.
Fig.3, on page 53, shows the drain-to-source waveforms of the four power FETs in the active clamp forward circuit. A complete switching period can be broken down into six individual switching periods to simplify the description of a full cycle. The transitions between each switching period are labeled as t0 to t6 in the figure.
Starting at time t0, the main switch (Q1) is on and the primary winding on the transformer sees the full input voltage. The reflected voltage on the transformer secondary turns on the forward MOSFET (Q3), while at the same time turning off the freewheeling MOSFET (Q4). Current is being supplied to the load from the input source, and energy is being stored in the magnetizing and leakage inductance of the transformer. With Q2 off, the clamp capacitor is fully charged and disconnected from the circuit during this portion of the switching cycle. Fig. 4 shows the current flow during this portion of the cycle.
At time t1, the Q1 turns off. As the energy stored in the leakage and magnetizing inductances begins to charge the drain-source capacitance of Q1, the voltage across the primary of the transformer begins to drop. The drain-source capacitance of Q1 continues to charge until its voltage is equal to the voltage of the fully charged clamp capacitor. As the drain-source capacitance of Q1 is charged, the voltage across the transformer primary decreases, and eventually reverses polarity.
The voltage on the transformer secondary eventually decreases below the turn-off threshold of Q3, which allows Q3 to turn off and Q4 to turn on. During this transition, cross-conduction between Q3 and Q4 is prevented by the fact that the drain of Q3 is connected to the gate of Q4. Q4 cannot turn on until Q3 has completely turned off.
At time t2, the drain-source voltage of Q1 is equal to the voltage on the clamp capacitor, and the body diode of Q2 begins to conduct. The remainder of the leakage energy is dumped into the clamp capacitor during this portion of the switching cycle. The magnetizing current decreases as its energy is transferred into the clamp capacitor. The transformer primary voltage remains at a negative potential, as the core is reset. The negative voltage on the transformer secondary ensures that Q3 is off, while Q4 is on. The inductor current is freewheeling through Q4 during this portion of the switching cycle.
At time t3, Q2 is switched on. Since the body diode of Q2 conducts before Q2 is switched on, Q2 turns on with a zero-voltage transition. This reduces switching losses in Q2. After Q2 is turned on, the magnetizing current continues to decrease to zero amps and eventually reverses. The control IC must turn Q2 on before the magnetizing current attempts to reverse directions. The exact delay before Q2 is turned on is controlled by the control IC. During this portion of the switching cycle, the inductor current continues to freewheel through Q4.
At time t4, the magnetizing current reverses directions, as some energy is removed from the clamp capacitor. The inductor current continues to circulate through Q4, which remains on.
Towards the end of the switching cycle, at time t5, Q2 turns off. The controller's oscillator determines the timing of this action. When Q2 turns off, the clamp capacitor is removed from the circuit, and remains charged. The drain-source capacitance of Q1 charges up until its voltage is equal to the input voltage, and the voltage drop across the transformer primary drops to zero volts. At the same time, the transformer secondary voltage drops low enough to turn off Q4.
A new switching cycle begins at time t6, where Q1 turns on again. The UCC3580 provides a programmable delay between the turn off of Q2 and the turn on of Q1. This delay ensures that the drain-source capacitance of Q1 has been discharged to the input voltage, which reduces the turn on losses in Q1. Cross-conduction between Q3 and Q4 is prevented by the fact that the drain of Q4 is connected to the gate of Q3. Q3 cannot turn on until Q4 has turned off.
A 100W, 3.3V forward converter employing active clamp control with self-driven synchronous rectifiers is shown in Fig. 5, on page 53. While the schematic of this design is as shown in Figs. 6 and 7, on page 54. It uses the UCC3580-1 controller to provide control of the switching regulator. Here, the main switch is an N-channel FET (Q1), and P-channel FET (Q2) is utilized as the clamp FET. The controller provides the drive for both MOSFETs.
This active-clamp design uses two bias voltages, one for the primary side and one on the secondary side. The output inductor (L1) contains an auxiliary winding, which is used to produce the primary-side bias, while the circuit composed of C100, C101, D100, D101, Q100, and R101 produces the secondary-side bias.
Breaking the feedback path with R19 and R20 provides remote sensing. By connecting the remote sense terminals directly to the desired regulation point, the converter compensates for any voltage drops between the output of the supply and the load. If remote sensing is not required, R19 and R20 may be shorted, in which case regulation will be provided directly at the converter output filter.
The feedback network is typical of most isolated forward converters. The TLV431 (U2) incorporates a voltage reference and transconductance error amplifier into one package. Providing type III compensation around U2 compensates the voltage-mode converter. The shutdown pin of the UCC3580-1 provides over-current protection. The current is sensed at the source of the main power FET by a current sense resistor composed of R2 and R103. Resistor R16 lowers the shutdown threshold voltage, which further improves the converter's efficiency.
The controller uses a built-in input under-voltage protection, while the comparator circuit of U4 provides input overvoltage protection. Both the input overvoltage and input under-voltage circuits provide hysteresis. The circuit containing U5 and U6 provides the output overvoltage protection.
The circuit operates from input voltages between 36V and 75V, at load currents up to 30A. The output voltages typically vary by only 4mV (0.1%) over the entire line and load range, see Fig. 8. The output ripple voltage of this design is kept below 26mV (0.8%) over line and load, as you can see in Fig. 9. Most importantly, this design is more than 90% efficient for nearly the entire operating range, as depicted in Fig. 10. At these power levels, obtaining an efficiency better than 90% is unlikely with a conventional RCD clamped converter.
Applying the Active Clamp Technique
By reducing losses in the main power switches and implementing self-driven synchronous rectifiers, an active clamped converter significantly improves the efficiency over conventional RCD clamped forward converters. Although not shown here, designers can also apply the active clamp technique to flyback converters. The increased efficiency of the active clamp forward converter allows the use of smaller power components, and consequently reduces the amount of board space consumed by the design.