For the past two decades, Insulated Gate Bipolar Transistors (IGBTs) and their associated silicon (Si)-based Free Wheeling Diodes (FWDs) have been the mainstay switching devices in 3-phase inverters. With recent advances in the ability to grow high-quality single-crystal silicon carbide (SiC) wafers, the time has come to recognize the true potential this technology has to offer. The advantages of SiC over its counterpart Si and gallium arsenide (GaAs) power-switching devices become apparent upon viewing their characteristic differences (Fig. 1).
The implementation of this technology in the field of motor control will come in two phases. The first phase will be the replacement of the Si PiN diode with a SiC Schottky Barrier Diode (SBD). The second phase will be the replacement of the IGBT with a SiC-equivalent transistor. This work is in progress and will permit further switching-loss reduction, leading to higher density power modules that can operate at significantly higher junction temperatures.
Schottky diodes are inherently capable of high-speed switching (< 50 nS), but previously have been based on Si technology, and thus limited to practical applications of < 200-V breakdown voltage due to the moderate field strength of Si. However, with SiC, the breakdown field strength is approximately a factor of 10 greater than Si. Therefore, the rated voltages can be a factor of 10 greater, permitting the use of these efficient high-speed devices in high-voltage inverters.
The SiC SBD's greatest advantage over its counterparts in high-power switching applications is its characteristic zero reverse recovery current, even at the highest junction temperature operation allowable. The SiC SBD, being a majority carrier, does not have any stored minority carriers, resulting in its minimal reverse recovery charge (Qrr) during turn-off. However, a small amount of displacement current is required to charge the Schottky junction capacitance, which is independent of current level, di/dt and most importantly, temperature. The junction charging current is dependent on dv/dt. Si PiN diodes inherently have a larger Qrr that increases dramatically with temperature (typically a factor of 3 from 25°C to 150°C), thus dramatically increasing switching losses in both the FWD and the IGBT.
The following data was obtained from a pair of popular off-the-shelf 230-V, 1-hp Variable Frequency Drives (VFDs). This particular unit was chosen because it contains discrete TO-220 IGBT/FWD co-packs as opposed to a power module for its inverter section, which lends itself to easier oscilloscope probing of the VFD when operating under rated load conditions. The currents and voltages that will be displayed in the oscilloscope plots that follow were observed in the U-phase leg of the 3-phase inverter section (Fig. 2).
IRG4BC20KD, the original co-packs, were removed and replaced with two TO-220 devices. For the IGBT, an International Rectifier IRG4BC20K was used. These were coupled to HFA15TB60 Si PiN diodes in one drive and to Cree Inc.'s CSD10060 SiC SBDs in the other. For each set of plots, the drive was run at rated load (i.e., 4.2 A and a PWM frequency of 10 kHz). The first plot shows the U-phase lower IGBT turn-on switching waveforms for the VFD modified with the Si PiN FWDs (Fig. 3).
Parameters worth noting include:
Peak reverse recovery current (Irr)-4.8 A
Peak IGBT collector current (Ic)-10.5 A
Reverse recovery time (trr)-112 ns
Switching loss (Eon)-0.16 mJ/pulse.
Fig. 4 shows the IGBT turn-on switching waveforms for the VFD modified with the SiC FWDs.
Parameters worth noting include:
Peak reverse recovery current (Irr)-0.8 A
Peak IGBT collector current (Ic)-7.5 A
Reverse recovery time (trr)-20 ns
Switching loss (Eon)-0.069 mJ/pulse.
The most notable improvement between the two technologies is the difference in diode reverse recovery current. The SiC Diode reverse recovery current was a factor of 6 less. Reverse recovery time also decreases by almost a factor of 6. This reduction is also reflected in the peak collector current flowing through the IGBT. As the IGBT turns on, the diode reverse recovery current becomes additive, as can be seen by the 40% increase in peak collector current between the two plots. All these improvements lead to the 57% reduction in IGBT turn-on loss that was observed.
The switching comparisons during IGBT turn-off were observed next (Figs. 5 and 6). A summary of the percentage reduction in switching losses and other noted parameters between the two technologies is available in the table.
The switching losses were calculated using standard power loss calculations for 3-phase sinusoidal inverters. Data was taken from the scope plots and parameters from the respective components' data sheets.
The Thermal Difference
The reduction of the overall losses in the inverter section presents many beneficial options to the VFD design engineer. The switch to SiC FWDs obviously presents an opportunity for a reduction in heat sinking and convection (natural or forced air/liquid) cooling requirements. This improvement impacts both cost (reduced heatsink sizing and/or fan sizing) and a reduction in physical size of the drive. The quest to reduce size while maintaining equivalent or more power output (increased power density) will always be a primary goal for any new VFD development program.
The difference in heatsink temperature was recorded between the two VFDs while running full rated load (4.2 A) with 10-kHz PWM frequency at a 40°C ambient (Fig. 7). The thermocouples were mounted in the heatsink directly beneath the bonding surface with the lower IGBT in the phase leg being monitored.
As expected from previewing the switching waveform and inverter loss data, the switch to SiC diodes results in a cooler running inverter. The delta from ambient in this case decreased from 14.2°C to 12°C, which is equivalent to a 15.5% reduction.
The VFDs used to gather the data above were selected because of the presence of discrete TO-220 packages in the inverter section. This enables easier probing for data collection during running conditions. By far, the majority of VFDs on the market today utilize power module technology. Typically, in drives of this size, the module contains both the inverter and rectification sections, and if need be, a seventh IGBT for dynamic braking. The power module is mounted to a heatsink, and typically the thermal interface is some form of silicon grease with a thermal resistance between module base-plate and heat sink, somewhere in the region of 0.1°C/W to 0.2°C/W.
Because the IGBTs used in this VFD have their metal tabs floating at collector voltage, the thermal interface is actually a heatsink type of isolation material, with all the TO-220 packs being torqued down to the heatsink by means of a small metal bar across all six of them. The metal bar is then fastened to the heatsink by means of a screw at either end. This is not an ideal interface and yields a 6.87°C/W thermal resistance after measuring the IGBT case temperatures.
At 40°C ambient, with the Si FWDs, the IGBT junction temperature calculated out to 77.5°C and the FWD junction temperature to 70.8°C. With SiC FWDs, the IGBT junction temperature dropped to 70.4°C and the FWD junction temperature to 64.5°C. This represents approximately a 9% decrease in junction temperatures for both the IGBT and FWD. Note that this IGBT percentage difference in junction temperature will actually increase as steady-state junction temperatures increase, by virtue of the characteristic increase in reverse recovery current inherent with Si devices. Looking at the delta in IGBT junction temperature with respect to ambient, the SiC FWDs present an 18.5% reduction.
For VFDs that operate at steady-state junction temperatures close to 120°C with Si FWDs, this reduction represents a significant improvement. It not only gives the extra headroom for increased power output, but also offers greater overload capability within the same package. Another important benefit is the fact that every 10°C decrease in steady-state junction temperature represents a doubling of the device's life expectancy.
PWM Carrier Frequency
PWM carrier frequency is another VFD variable that can benefit from these reduced inverter losses. Look at the example again from the perspective of increasing PWM frequency but maintaining the same power output. With the switch to SiC FWDs, the PWM frequency can be increased to 22.24 kHz before realizing the same inverter losses with Si FWDs. This is a significant increase and could be especially helpful when dealing with more specific applications where the audible carrier frequency noise is a concern. The VFD carrier frequency is now outside the human hearing range.
Another option the design engineer can take advantage of with these improvements is to investigate the possible increase in power output of the VFD package by switching to SiC FWDs and maintaining the same PWM frequency. In this example, the power output of the VFD can increase approximately 20% while maintaining a 10-kHz PWM frequency to match the equivalent inverter losses seen with the Si FWDs. However, note that when evaluating increased power output, the dc bus capacitors, dc link choke and input bridge may or may not need to be modified to handle the increase, depending on design margin. This needs to be evaluated by the design engineer on a case-by-case basis.
Since CE approval became mandatory in 1996 for all electrical equipment to be sold in Europe, EMI reduction to qualify for CE approval has become an integral part of every new VFD design program. Apart from passing several immunity tests, CE approval for VFDs also requires passing conducted emissions and radiated emissions, as outlined in the Industrial Generic Standard EN61800-3 guidelines for Adjustable Speed Electrical Power Drive Systems.
For the VFDs used in this article, an external RFI filter is required to comply with conducted emissions guidelines. The filter was not available during testing, so the focus here is on the radiated emissions difference between the VFD with Si and SiC FWDs. The units were tested in a certified indoor 10-meter anechoic chamber with 30 meters of shielded output cable driving an unloaded 1-hp motor. Peak data obtained from scans with both VFDs is shown in Fig. 8.
The improvement in radiated emissions is instantly apparent. The main reason for this reduction is again due to the inherent zero reverse recovery property of the SiC devices. As manufacturers of Si PiN diodes try and improve switching times, the reverse recovery currents in these devices become more “snappy.” The increased di/dt also can contribute to an increase in the peak voltage across the diode at recovery, adding to the VFD radiated EMI levels. The elimination of the recovery current in the SiC devices leads directly to the improved emissions spectrum observed.
In the 30-MHz to 40-MHz region, which is typically the noisiest realm with VFDs, the SiC diodes yield a 7 dB reduction at the highest peak around 34 MHz. Furthermore, there is a general 3-dB to 7-dB peak reduction across the 50-MHz to 200-MHz spectrum. This represents a significant improvement, especially to the VFD design engineer who spends many hours trying to find every last dB reduction possible in order to fall under the required limits.
Until now, the primary market for these higher voltage SiC SBDs has been predominantly in the switch-mode power supply and power factor correction markets. SiC enables the creation of near-perfect high-voltage diodes, and the advantages offered by this unique technology open up many new possibilities and opportunities in the motor drive and hybrid electric vehicle markets. The benefits realized by simply replacing the Si FWD with a SiC equivalent (thermal, EMI, frequency and reliability) are compelling.
Hodge, Stuart Jr.,“SiC Schottky Diodes in Power Factor Correction,” Cree Inc., http://powerelectronics.com/mag/power_sic_schottky_diodes_3/.
Wright — Patterson AFB, OH, “Silicon Carbide Schottky Diodes Improve Efficiency,” http://www.afrlhorizons.com/Briefs/Dec02/PR0205.html.
Singh, Ranbir, and Richmond, James, “SiC Power Schottky Diodes in Power Factor Correction Circuits,” Cree Inc., http://scdms05:8000/ftp/pub/CPWR-AN01.pdf.
|Parameter||Units||Si Pin||SiC SBD||Percent Reduction|
|Peak reverse recovery current Irr||A||4.8||0.8||83%|
|Reverse recovery time trr||nS||112||20 *||82%|
|Recovery charge Qrr||nC||241||28||88%|
|Switching loss per IGBT||W||1.2||0.7||41%|
|Switching loss per FWD||W||0.3||0.01||96%|
|Total switching loss per IGBT/FWD**||W||1.5||0.71||53%|
|* SiC trr is only due to junction capacitive charging.|
|** Assuming a 0.8 output power factor.|